In the text describing the prior art and the features of the present invention, the following abbreviations are used:
CAP Carrierless amplitude and phase modulation
DFE Decision-feedback equalizer
FFE Feedforward equalizer
LMS Least-mean-square
PAM Pulse amplitude modulation
QAM Quadrature amplitude modulation
RX Receiver
TX Transmitter
TML Tomlinson-Harashima precoder
Copper wirelines serving subscriber connections will be used for data transmission at increasingly higher speeds. Plausibly the transmission speeds will be extended to tens of megabits per second and higher. Herein, the bandwidth of the modulated signal will be in the order of 10 MHz.
When the modulation bandwidth reaches the megahertz range, different kinds of radio-frequency interference appear problematic particularly when overhead cables are used for subscriber connections. The modem receiver (RX) may also be subject to interference from radio stations emitting broadcast signals and radio-amateur stations, whose radio-frequency transmissions may be captured by the overhead cable so as to cause a so-called transverse interference signal component in the twisted pair cable and thus reach the receiver of the modem connection in an interference-causing manner. In a converse manner, the modem connection may disturb radio station listeners or reception at a radio-amateur station, because the power spectrum of the modulated signal traveling along the pair cable contains energy at the radio-frequency bands mentioned above and a portion of this energy is radiated to the environment.
The radio-frequency interference caused by the transmitter (TX) of the modem connection can be eliminated by way of filtering the power spectrum of the modem output signal free from emissions at the frequency bands coinciding with those subject to interference. The bandstop filters required herein may be implemented using analog, digital or both of these techniques. The radio-frequency interference coupled over the cable to the modem receive circuit may be filtered away by providing the receiver (RX) with suitable bandstop filters that may be implemented using analog, digital or both of these techniques, respectively.
In fast modems, signal processing generally is accomplished digitally, whereby the receive signal is first converted into digital format by an analog-to-digital converter (AD converter). Inasmuch the level of the radio-frequency interference emission may be high as compared with the received data signal, it is advantageous to in the optimum utilization of the dynamic range of the AD converter to perform the elimination of the radio-frequency interference at least up to a certain degree by means of bandstop filters that are realized with the help of the analog techniques and placed preceding the AD conversion.
On a modem connection, the signal transmission path (cable, line transformers, filters, etc.) cause amplitude and phase distortion on the signal that is generally compensated for by means of equalizers that are located in the receiver, the transmitter or partially in both. Bandstop filters used in the receiver, the transmitter or in both of these communicating units are particularly problematic as to the signal distortion, because the filters form a signal-eliminating stop band (or stop bands) on the signal spectrum, thus blocking or essentially weakening the transmission of certain frequency components of the data signal, too. In the discussion to follow, the bandstop filters located in the receiver, the transmitter or partially in both are considered to form a portion of the transmission channel.
An equalization technique of channel distortion commonly used in modems based on a linear modulation system (PAM, QAM, CAP) is to employ an adaptive linear equalizer (FFE) and an adaptive feedback equalizer. If the feedback equalizer is located in the receiver, it is called a decision-feedback equalizer (DFE) and, respectively, if located in the transmitter, it is known as a Tomlinson-Harashima precoder (TML). The system may also be provided with both the DFE and the TML. Furthermore, the linear equalizer may be located in the receiver, the transmitter or a part thereof in the transmitter and another part in the receiver.
In the following, a digital communications channel is examined in terms of the training phase of its adaptive equalizers. The line code used on the channel may be either a quadrature-amplitude modulation (QAM) or a carrierless amplitude and phase modulation (CAP). In FIG. 1 is shown a model for a system implemented using conventional techniques, wherein the receiver is provided with an adaptive linear equalizer (FFE) and an adaptive decision-feedback equalizer (DFE) (cf. Lee & Messerschmitt). The fixed filters and modulation schemes are included in the channel noise model (CHN). The outgoing bit stream is coded into symbols (S) that are sent through the channel 2. In the receiver, the output signal of the channel 2 is processed by equalizers (FFE and DFE), and the decisions on symbols (S′) are made from the equalized signal. The decision resulting in the resolved symbol (S′) is also called the estimated received symbol. The transmitted bit stream is recovered on the basis of the estimated received symbols (S′) resolved by the receiver (RX). Both adaptive equalizers are adapted to the characteristics of the channel 2 during the training period carried out when a connection is being established. The equalizers are also continually adjusted during the period of data transmission in order to compensate for possible changes in the channel 2. The equalizers are adapted and controlled on the basis of the detection error (e) of the receive signal with the help of a least-mean-squares (LMS) algorithm.
In FIG. 2 is shown another system according to the prior art (cf. Lee & Messerschmitt). The receiver has an adaptive linear equalizer (FFE), while the transmitter has a feedback equalizer (of the TML type). During the training period, also this system operates in the same fashion as that illustrated in FIG. 1 using a linear equalizer (FFE) and a decision-feedback equalizer (DFE). At the end of the training period, the tap-weight values of the decision-feedback equalizer (DFE) are transmitted over an upstream auxiliary channel to the transmitter, wherein they are utilized in the configuration of a Tomlinson-Harashima precoder (TML).
The methods generally used for training an adaptive channel equalizer to the characteristics of a transmission channel are blind training, which takes place from data, and forced training. In blind training, the transmitter (TX) sends during the training period a similar signal as that of a normal data transmission state and the receiver (RX) has no a priori information on the sent symbols. Hence, blind training is based on the statistical properties of the receive signal, wherefrom the receiver can adjust the equalizers into a correct state (e.g., assuming that each symbol of the symbol constellation occurs at an equal statistical probability). Generally, it is necessary to use a dedicated training detector during the training period in the same fashion as a circle detector is used in certain types of voice-frequency modems, for instance. FIG. 3 shows the timing diagram of the functional steps of the timing diagram of blind training. In forced training, the receiver (RX) has a priori information on the data (i.e., the training sequence) being sent by the transmitter (TX) during the training period. The equalizer adjustment is based on the difference values between the a priori known training sequence and the received signal. A problem hampering forced training is that the transmitter (TX) must submit the receiver (RX) signaling information on the start instant of the training sequence. Under the circumstances of practical channels, the transmission of such a start signal with a sufficiently high timing accuracy is often difficult. FIG. 4 shows a timing diagram of the functional steps of forced training.
In blind training the stop bands placed on the signal spectrum cause problems. To generate such stop bands, the transfer function of the communications channel must have one or more nulls that fall within the frequency spectrum of the data signal. These spectral nulls deteriorate the transmission of certain frequency components of the data signal so much that in terms of channel equalization, it is more correct to discuss the nulls of the transmission channel frequency spectrum that entirely prevent the transmission of the data signal at its frequency components falling on the spectral nulls of the channel. A linear equalizer (FFE) is incapable of fully compensating for the distortion caused by the spectral nulls of the channel, because a complete compensation of a spectral null in the channel transfer function would require infinite gain in the transfer function of the linear equalizer (FFE) at the frequency of each spectral null of the transmission channel. In other words, when there are spectral nulls in the transfer function of the channel and only a linear equalizer (FFE) is available, the equalizer output will inevitably bear some degree of intersymbol interference (ISI). The degree of intersymbol interference in proportion to the signal energy at the output of the linear equalizer (FFE) is dependent on such factors as the number, frequencies and Q-factors of the spectral nulls. Intersymbol interference complicates correct decision-making on received symbols in the same manner as noise. On the other hand, a sufficiently high proportion of the symbol decisions should be correct in order to permit the decision-feedback equalizer (DFE) to adapt so as to perform the compensation of channel distortion. Hence, blind training fails if the channel transfer function has such spectral nulls that a linear equalizer (FFE) alone is incapable of giving a sufficiently good symbol error rate for the adaptation of the decision-feedback equalizer (DFE).